1. Field of the Invention
The present invention is directed to a power supply that rectifies and smooths an AC voltage and provides a high frequency electric power through an inverter. In particular, the present invention relates to "charge pump" electronic ballast circuits for discharge lamps (e.g., fluorescent lamps). More specifically, the invention relates to the structure and operation of "charge pump" electronic ballast circuits having reduced DC bus voltages and voltage stresses at light loads, and incorporates improvements in input power factor, in reduced total harmonic distortion (THD) of the input current, and in reduced crest factor of the lamp current. Various deficiencies are overcome in a lamp preheat; start-up and normal lighting operations of the discharge lamps.
2. Discussion of Background And Other Information
In a prior power supply utilizing a rectifier for full-wave rectification of an AC voltage and a smoothing capacitor for smoothing the resulting DC voltage to provide a DC voltage, the AC current will not flow into the rectifier during a period in which an output voltage of the rectifier is less than a terminal voltage across the smoothing capacitor, thereby leaving a period of time (e.g. an off period) in which there is no current flow. The presence of the off-period brings about a lag between the input current and the voltage to thereby lower an input power factor and to produce a noise due to the input current distortion.
To overcome this problem, it is necessary to continue supplying the input current irrespective of the output voltage level from the rectifier. To this end, it has been proposed to feed back the high frequency output from an inverter supplied with the smoothed DC voltage which is obtained by rectifying and smoothing the input AC voltage, in order to interrupt the input current from the AC voltage source at a frequency sufficiently higher than the frequency of the AC voltage source, thereby avoiding the appearance of the off-period in the input AC current continuing over about one cycle of the input AC current.
One example for achieving the above proposal is shown in FIG. 17. This example has the configuration disclosed in Japanese Patent Early Publication No. HEI 7-147778. This configuration includes a rectifier RE that provides full-wave rectification of the input AC current from the AC voltage source, such as an AC mains, and a smoothing capacitor C1 for smoothing the output from the rectifier RE to produce a DC voltage. The resulting DC voltage is converted by an inverter 1 into a high frequency electric power, which is then applied to a discharge lamp La, i.e., a load through an output transformer T1. The output transformer T1 includes a pair of pre-heating windings np, each connected across each filament F1 and F2 of the discharge lamp La for supplying a pre-heating current to the filaments F1 and F2. Also included in the output transformer T1 is a feedback winding n13 which is connected in series with a capacitor C2 to form a series circuit of the feedback winding n13 and capacitor C2. This circuit is inserted between the input terminals of the rectifier RE. An inductor L2 is connected between the AC voltage source and the rectifier RE to form a resonant circuit with capacitor C2. Connected between the inductor L2 and the AC voltage source AC is a line filter F. The resonant circuit of inductor L2 and capacitor C2 is designed to have a resonant frequency that is nearly equal to an output frequency of the inverter 1.
An equivalent circuit of the above configuration is shown in FIG. 18, in which the rectifier RE is connected across the AC voltage source through line filter F (only capacitor CF is shown in the figure) and the resonant circuit 2. The smoothing capacitor C1 is connected to smooth the output of the rectifier RE. The inverter 1, deriving its input voltage from the capacitor C1, is shown as a load Z. The resonant circuit 2 comprises capacitor C2 connected in series with a high frequency voltage source VR and inductor L2 connected between the line filter F and the rectifier RE. The series circuit of capacitor C2 and the high frequency voltage source VR is connected across the input terminals of the rectifier RE. The high frequency voltage source VR corresponds to feedback winding n13.
Considering a positive half-cycle of the AC voltage source, a high frequency voltage induced at feedback winding n13 alternates so that two periods alternate with each other; one period in which capacitor C2 is charged by a current flowing through AC voltage source--inductor L2--capacitor C2--high frequency voltage source VR and the other period in which an output voltage of the high frequency voltage source VR is applied across capacitor C2 and then to rectifier RE. That is, within one cycle of the high frequency voltage source VR, there appears a certain period in which AC voltage source flows a charge current to capacitor C2, so that it is possible to flow an input current from the AC voltage source at a frequency sufficiently higher than that of the AC voltage source. With the use of a line filter F (expressed by capacitor CF) provided between the AC voltage source and resonant circuit 2, the waveform of the input current from the AC voltage source becomes an envelop of the waveform of the input current supplied to the resonant circuit 2, so that the input current will flow continuously from the AC voltage source at a level generally proportional to the voltage of the AC voltage source. Thus, a high power factor and a low input current distortion can be achieved. Although only the positive half-cycle is described in the above, a negative half-cycle operates in the same manner.
When the above operation is done, an input voltage to the rectifier RE varies, as shown in FIG. 19, between an upper extreme (E+E0) and a lower extreme (--E--E0), wherein E is the voltage of the AC voltage source and E0 varies with an induced voltage across the feedback winding n13 to be nearly twice as large as the induced voltage. Thus, there appears a period in which the voltage exceeds the peak voltage of the AC voltage source.
As shown in FIG. 20, a scheme was proposed in Japanese Patent Early Publication No. HEI 5-38161 to avoid the off-period of the input current by feeding back a portion of the high frequency output from the inverter 1 to the input side. This configuration does not include the resonant circuit 2 between the filter F and rectifier RE, but includes means to feed back a portion of the high frequency output from the inverter 1 to the input side of the inverter after making full rectification through the rectifier RE. A parallel circuit of a feedback capacitor C4 and a diode D3 is inserted between the rectifier RE and a smoothing capacitor C1. The polarity of diode D3 is selected in order to flow a charging current from the rectifier RE to the smoothing capacitor C1.
The inverter 1 comprises a series connected pair of switching elements Q1 and Q2 connected in parallel with the smoothing capacitor C1. A series circuit of a DC blocking capacitor C0, a load, e.g., discharge lamp La, and the inductor L3 is inserted between the positive output terminal of the rectifier RE and the connection point of the switching elements Q1 and Q2. The switching elements Q1 and Q2 are MOSFET transfers which are controlled by a switching controller CN to alternately turn on and off without being caused to turn on simultaneously. It is noted that the switching elements Q1 and Q2 include internal parasitic diodes D1 and D2. The discharge lamp La includes filaments F1 and F2 between which a pre-heating capacitor Cp is connected.
The inverter 1 operates as follows: it is noted that a current flows through filaments F1 and F2 and capacitor Cp to preheat the filaments F1 and F2 before the discharge lamp La is turned on, and that the capacitor Cp is disconnected from the circuit upon turn-on of the discharge lamp La. In the following explanation, the discharge lamp La and the capacitor Cp are regarded as the load circuit.
The inverter 1 operates differently depending upon the relation between the output voltage of the rectifier RE and the voltage developed across the smoothing capacitor C1. Considering a period in which the output voltage of the rectifier RE which exceeds the voltage of the smoothing capacitor C1, the diode D3 is made conductive to flow the charging current from the rectifier RE to the smoothing capacitor C1. When the switching element Q2 is made conductive in this period, a current flows from the rectifier RE through capacitor C0--load circuit--inductor L3--switching element Q2. Subsequently, when the switching element Q2 is turned off, energy stored in the inductor L3 is released through a path of parasitic diode D1--smoothing capacitor C1--rectifier RE--capacitor C0--load circuit. Thereafter, when the switching element Q1 is made conductive, capacitor C0 is discharged to flow a current though a path of diode D3--switching element Q1--inductor L3--load circuit. Upon subsequent turn-off of the switching element Q1, energy stored in inductor L3 is released through a path of the load circuit--capacitor C0--diode D3--smoothing capacitor C1--parasitic diode D2. That is, during this period, in which rectifier RE provides an output voltage higher than the voltage of the smoothing capacitor C1, the diode D3 becomes conductive to provide no current flow through capacitor C4 and the load circuit receives a high frequency alternate current in accordance with the above operations.
On the other hand, during a period in which the output voltage of the rectifier RE is less than the voltage of the smoothing capacitor C1, diode D3 is made not conductive and capacitor C4 becomes operative. This condition is expressed by V1.sup.3 Vin and V1+V4=Vin, wherein V1 is the voltage across smoothing capacitor C1, V4 is the voltage across capacitor C4, and Vin is the output voltage of rectifier RE. V4 .English Pound. 0 means that capacitor C4 acts to absorb the difference between the voltage across the smoothing capacitor C1 and the output voltage of the rectifier RE. With this result, it is possible to flow the current from the rectifier RE to the inverter 1 even during the period in which the output voltage of the rectifier RE is less than the voltage across the smoothing capacitor C1, thereby elongating the period of flowing current from the AC voltage source than without the capacitor C4, and therefore reducing the off-period of flowing no current as well as reducing input current distortion.
The operation of the circuit shown in FIG. 20 will be explained in more detail. When the switching element Q2 is made conductive, current flows from smoothing capacitor C1 through a path of capacitor C4--capacitor C0--load circuit--inductor L3--switching element Q2. At the same time, current flows from rectifier RE through a path of capacitor C0--load circuit--inductor L3--switching element Q2. Upon turn-off of the switching element Q2, energy stored in inductor L3 is released through a path of parasitic diode D1--capacitor C4--capacitor C0--load circuit. When the switching element Q1 is turned on, current flows from capacitor C4 through switching element Q1--inductor L3--load circuit--capacitor C0. Upon subsequent turn-off of switching element Q1, energy stored in inductor L3 is released through a path of load circuit--capacitor C0--capacitor C4--smoothing capacitor C1--parasitic diode D2.
As apparent from the above explained operations, during the period in which the output voltage of the rectifier RE is less than the voltage across the smoothing capacitor C1, the capacitor C4 repeats being charged and discharged in response to the turn-on and turn-off of the switching elements Q1 and Q2. Since the charging and discharging are accompanied with a condition where capacitor C4 is charged by the energy from the inverter 1 and a condition where capacitor C4 is discharged to charge smoothing capacitor C1, capacitor C4 can be said to have a feedback function of delivering a portion of the output from the inverter 1 to charge smoothing capacitor C1.
FIG. 21 shows a configuration which is disclosed in Japanese Patent Early Publication (KOKAI) No. HEI 4-193067 to feed back a portion of the high frequency output of the inverter 1 to the input side thereof. Like in the above described configuration, this configuration utilizes no resonant circuit 2 and is arranged to feed back a portion of the high frequency output of the inverter 1 to the input side of the inverter after fill-wave rectification by the rectifier RE. As compared with the circuit of FIG. 20, a difference is seen in that capacitor C4 for feeding back a portion of the high frequency output of the inverter 1 is inserted between a positive terminal of the rectifier RE and a DC blocking capacitor C0, rather than being connected in parallel with the diode D3, and that a load circuit is connected in series with the capacitor C4 across the rectifier RE.
Inverter 1 is generally of the similar configuration as shown in FIG. 20, and comprises a pair of switching elements Q1 and Q2 connected across a smoothing capacitor C1. The DC blocking capacitor C0 is connected in series with an inductor L3 and feedback capacitor C4 between a positive terminal of rectifier RE and a connection point of the switching elements Q1 and Q2. The switching elements Q1 and Q2 are MOSFETs and are controlled by a switching control circuit (not shown) to alternately turn on and off without being caused to turn on simultaneously. The switching elements Q1 and Q2 include internal parasitic diodes D1 and D2. The discharge lamp La includes filaments F1 and F2 between which a pre-heating capacitor Cp is connected.
The inverter 1 operates as follows: when the switching element Q1 is made conductive, current flows from smoothing capacitor C1 through a path of switching element Q1, inductor--inductor L3--capacitor C0--load circuit Upon turn-off of the switching element Q1, energy stored in inductor L3 is released through a path of parasitic capacitor C0--load circuit--parasitic diode D2. Thereafter, when the switching element Q2 is turned on, current flows from capacitor C4 through a path of inductor L3--switching element Q2--load circuit. Upon subsequent turn-off of switching element Q2, energy stored in the inductor L3 is released through a path of parasitic diode D1--smoothing capacitor C1--load circuit--capacitor C0.
Since current will flow from capacitor C4 to inductor L3 when the switching element Q2 is conductive, as explained above, a current will also flow from rectifier RE through a path of capacitor C4--capacitor C0--inductor L3--switching element Q2. Capacitor C4 forms a resonant circuit with inductor L3 and switching elements Q1 and Q2 are turned on and off at a timing when nearly zero voltage is applied to the elements. Consequently, switching element Q1 is turned on at a polarity reversal of the current in the resonant circuit to flow current to the resonant circuit through a path of capacitor C4--diode D3--switching element Q1--inductor L3--capacitor C0, and through a path of capacitor C4--diode D3--smoothing capacitor C1--parasitic diode D2--inductor L3--capacitor C0.
That is, within one cycle in which the switching elements Q1 and Q2 are turned on and off, there appear two periods, one period in which current flows from the rectifier RE through the capacitor C4, and another period in which a portion of the high frequency output from the inverter 1 is fed back through the diode D3 to the smoothing capacitor C1 (the input side of the inverter 1), which enables the high frequency input current to continuously flow from the AC voltage source, thereby reducing the input current distortion. Also, since an envelop of the input current from the AC voltage source is made to be generally proportional to the input voltage, a high power factor is obtained.
The above described prior art configurations are designed to use the discharge lamp La as a load. Consequently, a control is made to pre-heat the filaments F1 and F2 for a predetermined period at the start of lamp lighting and subsequently to apply a starting voltage equal to about 3 or 4 times a normal lamp voltage required for keeping the lamp turned on, to start lighting the lamp, and thereafter to lower the voltage applied to the lamp for a stable lighting of the lamp. For example, with the circuit of FIG. 20, in the pre-heating period lasting over the predetermined period from the connection to the power source, the switching elements Q1 and Q2 are turned on and off at a frequency higher than a resonant frequency of a resonant circuit (including the inductor L3 and the load circuit) so as to flow current through the pre-heating capacitor Cp for pre-heating the filaments F1 and F2. Thereafter, the switching frequency approximates the resonant frequency to apply the resulting starting voltage to the discharge lamp La. Thus, the voltage is 3 or 4 times greater than the normal lighting voltage applied between the filaments F1 and F2, to start lighting the discharge lamp La.
In addition, lighting currently consumes up to approximately 25% of the total electrical energy used today. In the U.S. and throughout the world, government regulatory agencies have required the use of electronic ballast devices to save energy and to improve power quality. As demand for high frequency electronic ballast has grown rapidly in recent years, a number of innovative topographies have emerged. Among them, the "charge pump" electronic ballasts have gained popularity for use with discharge lamps due to their simplicity and low cost.
Since the electronic ballast is an ac/ac power processor converts the line frequency ac power into high frequency ac power to light the lamp, the ballast circuit consists of two stages: ac/dc rectification with a power factor correction (PFC) 100 and dc/ac inversion 101 (as shown in FIG. 22). FIG. 23 shows an example of a conventional electronic ballast: a PFC boost converter 102 followed by a parallel resonant inverter 103. Since there are two controls available in this circuit, good performance, such as good unity power factor and low crest factor of the lamp current, and good dimming of the light, etc., can be easily obtained. However, the two-stage approach requires two sets of power stages and control circuits. The cost of this electronic ballast is high. It is noted that a crest factor of the lamp current is defined as CF=I.sub.1a,pk /I.sub.1a,rms, and the peak value I.sub.1a,pk and the rms value I.sub.1a,rms of the lamp current are measured on the basis of one line cycle.
If the boost converter 102 is operated in a discontinuous current mode (DCM), some extent of PFC is naturally obtained. Therefore, two stages can be integrated into one stage. One example is shown in FIG. 24A. Two switches S1 and S2 are complementarily switched to drive the resonant inverter tank. At the same time, lower switch S1 implements the boost switch function, and anti-parallel diode D.sub.s2 of upper switch S2 functions as a diode in the boost converter. Fast diode D1 in series with boost inductor L.sub.in ensures the DCM operation of the boost inductor L.sub.in. In this topology, ripple across a dc bus is usually very small. Consequently, the crest factor of the lamp current can be low. Duty cycle control or frequency control can be adopted. With duty cycle control, the resonant tank current is sensed in order to turn on the MOSFET switch only when its body diode conducts. Otherwise, the reserve recovery current of the body diode may kill the MOSFET device. Under frequency control, the de bus voltage can increase significantly at light load operating conditions. An additional protection circuit is needed to prevent the switches from suffering the over voltage. The lower switch S1 in this circuit usually has a much larger current stress than the upper switch S2 because it has to take the sum of the boost inductor current and the resonant inverter tank current. Consequently, the size of the lower switch S1 must be larger than that of the upper switch S2. In order to reduce the THD of the input current, the dc bus voltage should be high enough. The voltage stress of the semiconductor devices can be high.
Another type of electronic ballast circuit, employing a charging capacitor and the high frequency source (either voltage source or current source) to implement PFC, was recently proposed. This type of ballast circuit is sometimes called "charge pump" circuit. FIG. 25A shows the principle diagram which employs the charging capacitor Cin and the high frequency ac voltage source (HFVS). By designing the dc bus voltage V.sub.dc to be higher than the input line voltage V.sub.g, the diodes D and DB will not conduct at the same time. The input current would then equal the positive charging current of Cin, which is regulated by V.sub.a, V.sub.g and V.sub.dc. If the charge variation of Cin (which is proportional to the variation of V.sub.c, as shown in FIG. 25B) follows the input voltage, the input average current will follow the input voltage, and a good input power factor can be obtained. One example of the "charge pump" circuit is shown in FIG. 6. Compared to the boost integrated circuit shown in FIG. 24A, this circuit replaces the boost inductor by a charging capacitor. It should be noted that the current stresses of the two switches in this circuit are then the same. Therefore, this type of circuit is potentially low-cost. However, the switches still suffer high voltage stress under light load conditions. Furthermore, due to an injection of line ripple through Cin, the crest factor of the lamp current and the THD of the line current can be high.
Due to the fact that the smoothing capacitor C1 becomes bulky with an increase in capacity, capacitor C1 is selected to have a capacity only sufficient to provide an adequate voltage input to the inverter 1 required to keep a stable operation of the inverter 1 (stable lighting of the discharge lamp La). That is, the smoothing capacitor C1 is selected to have a capacitance which provides a voltage less than when the smoothing capacitor C1 is charged directly from the output of the rectifier RE.
When the load of the inverter 1 becomes less, the input current to the inverter 1 is reduced, to thereby increase the voltage developed across the smoothing capacitor C1. This means that less electric power is consumed at the pre-heating and lamp starting, to thereby increase the voltage across the smoothing capacitor C1. Thus, the voltage across the smoothing capacitor C1 increases with a decrease in the load requirement. Particularly, at the time of starting the lamp which is a transition from the pre-heating to the lighting, switching frequency shifting is required to increase the voltage applied to the discharge lamp La, such that the voltage across smoothing capacitor C1 will increase largely with the circuit configuration of feeding back the output voltage of the inverter 1 to the input side.
When designing the circuit in due consideration of the increase in the voltage developed across the smoothing capacitor C1, a high dielectric strength is required to the smoothing capacitor C1 and also to the associated components of the inverter 1 supplied with its input voltage from the smoothing capacitor 1, which incurs a problem of increasing component costs.